This invention generally relates to optical communications, and in particular to a method and system for determining the amount of optical filtering present in an optical communications network and setting values for receiver characteristics according to the amount of optical filtering.
The backbone of point-to-point information transmission networks is a system of optically amplified dense wavelength division multiplex (DWDM) optical links. DWDM optical fiber transmission systems operating at channel rates of 40 Gb/s and higher are highly desirable because they potentially have greater fiber capacity and also have lower cost per transmitted bit compared to lower channel rate systems.
The modulation format of 40 Gb/s DWDM transmission systems is typically chosen to have high Optical Signal-to-Noise Ratio (OSNR) sensitivity. High OSNR sensitivity means that a low OSNR is sufficient to maintain a desired bit error ratio (BER) of the transmission or, equivalently, that the system is able to operate at a desired BER even in the presence of a high level of optical noise. In addition, modulation formats of 40 Gb/s DWDM transmission systems are typically chosen to be tolerant to optical filtering because existing systems sometimes include optical multiplexers and demultiplexers for 50 GHz channels spacing that limit the bandwidth. Also, existing systems sometimes include cascaded optical add-drop multiplexers.
Accordingly, Differential Phased Shift Keying (DPSK) has been considered for 40 Gb/s DWDM transmission systems, in part because DPSK transmission systems have excellent OSNR sensitivity. DPSK transmission systems using balanced direct detection receivers, which are sometimes referred to as differential receivers, have been shown to have an approximately 3 dB improvement of OSNR sensitivity compared to on-off keying systems, such as NRZ and PSBT systems. However, conventional DPSK transmission systems do not have good filter tolerance.
In a DPSK system, data is encoded onto a carrier wave by shifting the phase of the carrier wave. The amount of the phase shift may be selected based on the amount of data to be encoded with each phase shift. For example, DPSK is sometimes referred to as Differential Binary Phase Shift Keying (DBPSK). In DBPSK, the phase of the signal may be shifted in increments of 180° (i.e., by it radians) in order to encode a single bit of data (“1” or “0”) with each phase shift. In another example, in Differential Quadrature Phase Shift Keying (DQPSK), the phase of the signal may be shifted in increments of 90° (i.e., by π/2 radians) in order to encode two bits of data (e.g., “11” or “01”) with each phase shift.
The number of possible phase shifts is typically referred to as the number of “constellation points” of a modulation format. For example, DPSK has two constellation points, and DQPSK has four constellation points. Modulation formats using different number of constellation points (e.g., “m” constellation points) are also known, and are referred to generically as DmPSK formats.
If both the phase of the signal and the amplitude of the signal are used to encode the signal with the data, then the modulation format is called QAM (quadrature amplitude modulation) or m-QAM, where m is the number of constellation points.
A shift in the phase of the signal is referred to as transmitting a “symbol,” and the rate at which each symbol is transmitted is referred to as the “symbol rate.” As noted above, multiple bits of data may be encoded with each symbol. The rate at which bits are transmitted is referred to as the “bit rate.” Thus, the symbol rate in a DQPSK system is half the bit rate. For example, a DPSK system and a DQPSK each transmitting at the same symbol rate would evidence different bit rates—the DQPSK system would have a bit rate that is twice the bit rate of the DBPSK system.
Accordingly, a 43 Gb/s data rate in a DQPSK system corresponds to 21.5 Giga symbols per second. Thus, DQPSK transmission systems have a narrower spectral bandwidth, greater chromatic dispersion tolerance and greater tolerance with respect to polarization mode dispersion (PMD) compared to traditional formats and compared to DBPSK. However, DQPSK transmission systems have approximately 1.5-2 dB worse receiver sensitivity than DBPSK transmission systems. Furthermore, both the transmitter and the receiver are significantly more complex than a traditional DBPSK transmitter/receiver.
DBPSK and DQPSK can be of the non-return-to-zero (NRZ)-type or, if a return-to zero (RZ) pulse carver is added to the transmitter, may be of the RZ-type.
FIG. 1A is a block diagram describing an example of optical network 100 for transmitting, among other things, a DPSK optical signal.
A transmitter 102 may generate a DPSK optical signal 104. The transmitter 102 may include, for example, a light source such as a laser. A pulse carver may accept light from the light source and add a pulse to the light. The pulsed beam may have a phase which can be manipulated by one or more modulators in order to encode a data signal on the light. The manipulated light may be a DPSK optical signal 104.
The DPSK optical signal 104 may be combined with one or more other signals 106, such as on-off-keyed (OOK) signals, at a multiplexer 107. For example, the signals may be multiplexed using wavelength division multiplexing (WDM), and two neighboring signals may have relatively similar wavelengths. By multiplexing the signals 104, 106 together and/or filtering the signals using one or more optical filters 108, more information can be carried over a transmission line 109. The filters 108 may include, for example, multiplexers, demultiplexers, optical interleavers, optical add/drop filters, and wavelength-selective switches. The filters 108 may spectrally narrow the signal passing therethrough.
The combined optical signal carried on the transmission line 109 may be received at a receiver 110 for demodulating the combined optical signal. Prior to the receiver 110, a demultiplexer 111 may receive a multiplexed signal. The demultiplexer 111 may select one of the signals, for example the DQPSK signal 104. The demultiplexer 111 may select the signal, for example, by isolating a particular wavelength carrying the DPSK signal 104. Alternatively, the receiver 110 may include a demultiplexer 111 or selector for receiving an incoming modulated optical signal.
The receiver 110 may receive a source beams 113. The source beam 113 is received at an interferometer 116.
DPSK receivers typically use one or more optical demodulators that convert the phase modulation of the transmitted optical signal into amplitude modulated signals that can be detected with intensity receivers. Typically, optical demodulators are implemented as delay line interferometers (DLIs) 116 that split the optical signal into two parts, delay one part relative to the other by a differential delay Δt, and finally recombine the two parts to achieve constructive or destructive interference depending on the phase which is modulated onto the optical signal at the transmitter 102. Thus, the interferometer may interfere a DPSK optical signal with itself.
The optical demodulator converts the DPSK phase-modulated signal into an amplitude-modulated optical signal at one output and into the inverted amplitude-modulated optical signal at the other output. These signals are detected with a photodetector 120, which may consist (for example) of two high-speed detectors (see, e.g., FIG. 1B). The outputs of the detectors are electrically subtracted from each other, and after that the resultant electrical signal is sent to the data recovery circuits.
In operation, the interferometer 116 shifts the phase of the incoming signal. For example, in a DPSK system, the interferometers 116, 118 may shift the phase of the incoming signals relative to each other by π. The interferometer 116 is used to analyze and/or demodulate the incoming modulated optical signal 102, and provide an outputs to a detector 120. The interferometer 116 is described in more detail below with reference to FIGS. 1B-1D.
The interferometer may generate one or more optical inputs for a detector 120. For example, the interferometer 116 may generate a first optical input 117 and a second optical input 118 that are provided to a photodetector 120. The photodetector 120 may operate on the input optical signals and generate an electrical output signal 124.
In some embodiments, the detector 120 may be, for example, a balanced detector or an unbalanced detector.
FIG. 1B is a block diagram of a portion of the receiver 110 of FIG. 1A. In the receiver 110, the interferometer 116 and photodetector 120 cooperate to turn a first optical source beam 113 in the optical domain into a first output signal 124.
At the interferometer 116, the optical source beam 113 is split into a sample beam 128 and a reference beam 130. The sample beam 128 and reference beam 130 are processed to generate a first optical input 117 and a second optical input 118, which are respectively received by first and second detectors 132, 134 in the photodetector 120. The first and second detectors 132, 134 include a first output port 136 and a second output port 138, respectively, for providing outputs to an electronic device. The electronic device may be, for example, a differential detector that subtracts the first output 136 from the second output 138 in order to generate the electrical output signal 124.
FIG. 1C is an example of an interferometer, such as, for example, interferometer 116. The interferometer 116 may be, for example, a delay line interferometer (DLI) which receives one of the signal components (e.g., the first source beam 113) from the splitter 112. The interferometer 116 may be fabricated, for example, in gallium arsenide or lithium niobate, free-space optics (e.g., FIGS. 1C and 1D), fiber (e.g., FIGS. 1B, 2, 3) or PLC. Other examples of interferometers include Mach-Zehnder interferometers (MZIs).
The interferometer 116 may include a first splitter 142 for splitting the received first source beam 113 into two or more interferometer signal components 128, 130. The first interferometer signal component 128 is referred to as the sample beam, and is provided to a first mirror 148 along an optical path 144. Likewise, a reference beam 130 is supplied to a second mirror 150 along a second optical path 146. The optical paths 144, 146 may include an optical medium through which the signals travel. For example, the optical paths 144, 146 may include air or glass. The optical properties of the medium in the optical paths 144, 146 affect the amount of time that it takes the signals 128, 130 to travel in the optical paths 144, 146.
From the mirrors 148 and 150, the respective interferometer signal components 128 and 130 are provided to another splitter 152, where the signal is further split into a pair of signals (a first optical input 117 and a second optical input 118), which are received by two or more detectors 132, 134.
If the optical paths 144, 146 (or other optical paths not pictured) are identical in length and other properties, then the sample beam 128 and the reference beam 130 arrive at the detectors 134, 136 at the same time. However, by varying one or more of the optical paths 144, 146 with respect to the other, a time delay can be introduced, as shown in FIG. 1D.
As depicted in FIG. 1D, the interferometer 116 may be unbalanced in that the interferometer has a time delay 154 (often referred to by the symbol “τ”), which in some situations may be equal to the symbol period (e.g., 50 ps for a 20 Gsymbol/s line rate) of the data modulation rate, in one optical path 144 relative to that of the other optical path 146. The time delay 154 affects the time at which each respective beam 128, 130 is received at the detectors 132, 134.
Using binary phase shift keying, the phase of a signal may be shifted in two different ways (by 0 or π). Accordingly, each phase shift can encode a signal having a bit of information (e.g., “0” or “1”). The symbol rate refers to the rate at which these “symbols” are transmitted in the network (e.g., the number of symbol changes made to the transmission medium per second), while the symbol period refers to the amount of time that it takes for a single symbol to be transmitted. For example, if it takes 46.5 ps (i.e., 4.65×10−11 seconds) to transmit a single symbol, then the symbol period is 46.5 ps and the symbol rate is approximately 2.15×1010 symbols per second (or 21.5 Gsymbol/s).
Conventional interferometers include a time delay 154 in order to determine the amount that a particular signal has been phase shifted. Conventionally, the time delay 154 may be set to (for example) one symbol period in order to aid in the interpretation of the phase shifted signal. However, the time delay 154 may also be set to be larger or smaller than the symbol period, as discussed in U.S. patent application Ser. No. 12/906,554, entitled “Method And System For Deploying An Optical Demodulator Arrangement In A Communications Network” and filed Oct. 18, 2010, the contents of which are incorporated herein by reference.
In the “classical” implementation of DPSK receivers, the time delay 154 between the two arms of the interferometer is an integer number of the time symbol slots of the optical DPSK data signal: Δt=n T (where n=1, 2, . . . T; T=1/B is the symbol time slot; and B is the symbol bit-rate).
The time delay 154 may be introduced by making the optical path length of the two optical paths 144, 146 different, or may be introduced by varying the medium through which one of the signals 128, 130 travels, among other things. For ease of fabrication, the time delay 154 may be introduced by making the physical length of the interferometer's 116 optical path 144 longer than the physical length of the other optical path 146.
The interferometer 116 is respectively set to impart a relative phase shift 156 by the application of an appropriate voltage to electrodes on the shorter optical path 146. The amount of the phase shift 156 may be determined, for example, based on the modulation format. In the example of DQPSK, the relative phase shift 156 may be π/4 or −π/4. In the example of DPSK, the relative phase shift 156 may be π or 0. A more detailed description of the interferometers and time delay can be found in U.S. patent application Ser. No. 10/451,464, entitled “Optical Communications,” the contents of which are incorporated herein by reference.
Changing the amount of time delay 154 can change the Free Spectral Range (FSR) of the interferometer 116. The FSR relates to the spacing in optical frequency or wavelength between two successive reflected or transmitted optical intensity maxima or minima of, for example, an interferometer.
An FSR of an interferometer may be modified in accordance with a change in the optical bandwidth of the optical signal passing through the interferometer. Until recently, it was a common understanding that the best performance (best optical signal-to noise ratio OSNR sensitivity) is obtained when the time delay between the two arms of the interferometer Δt is exactly an integer number of the time symbol slots of the optical DPSK/DQPSK data signal, and the penalty increases rapidly (˜quadratically) when Δt deviates from its optimal value (see, for example, Peter J. Winzer and Hoon Kim, “Degradation in Balanced DPSK receivers”, IEEE PHOTONICS TECHNOLOGY LETTERS, vol. 15, no. 9, page 1282-1284, September 2003). In other words, according to conventional theory the optimum FSR (FSR=1/Δt) of the DLI equals 1/nT, and (in case of n=1) equals the symbol rate of the signal.
The performance of DPSK modulated optical networks considerably reduces when the signal is spectrally narrowed (for example, after going through optical multiplexers/demultiplexers, optical interleavers, optical add/drop filters, wavelength-selective switches or other filters 108, when the symbol rate B is comparable to the channel spacing in WDM transmission, etc). To improve performance of DPSK/DQPSK in such bandwidth-limited transmission, a concept of Partial DPSK (P-DPSK) was introduced: by making the time delay between the two arms of the delay interferometer Δt smaller than the symbol size T (or, equivalently, making the DLI FSR larger than the symbol rate: FSR>1/T), the performance of the optically-filtered DPSK was considerably improved (see, e.g., U.S. patent application Ser. No. 11/740,212, entitled “Partial DPSK (PDPSK) Transmission Systems” and filed on Apr. 25, 2007, the contents of which are incorporated herein by reference). It was shown that depending on the amount of the signal spectral filtering in the transmission system, an optimum FSR of the DLI exists, and this optimum FSR is different for different strength of optical filtering.
Nevertheless, in practical systems the receiver should be able to operate in conditions with different amount of the signal spectral filtering in the transmission line: for example, the combined optical bandwidth of systems with reconfigurable optical add/drop multiplexers (ROADMs) can change dramatically depending on the number of ROADMs in the system and the ROADMs settings. The receiver settings need to be optimized depending on the conditions (for example, noise, signal strength, optical filtering etc) in the transmission line. This requirement becomes more important in reconfigurable networks, where the transmission distance and the amount of optical filtering in the system may change dramatically during the operation.
A traditional approach to control the receivers is to provide a feedback from a forward correction (FEC) chip and adjust the receiver parameters to the minimum possible bit-error-ratio (BER) provided by the FEC chip. FIG. 2 depicts an example of a system employing FEC to adjust characteristics of the receiver.
In a FEC scheme, redundant data such as error correcting code (ECC) may be transmitted over the transmission line. The ECC may be predetermined and previously programmed into a detection unit located at the receiver. The ECC may be received at the receiver 110 together with a payload of the signal. A power supply 210 powers the balanced detector 120, which accepts the output of the photodetectors 132, 134 and subtracts them in a differencing unit 212 to interpret the signal. The differencing unit may be, for example, a transimpedance amplifier (TIA). Hence, the detector 120 may be a differential detector that outputs an analog electrical signal.
The resulting demodulated signal is provided to a clock and data recovery (CDR) unit 240, which is controlled by an FEC detection unit 220 and control unit 230. Note that if a limiting amplifier is used before the CDR, the amplifier is considered part of the CDR circuit because an amplitude decision is made by the amplifier.
The CDR circuit converts the receiving analog electrical signal into a digital electrical signal. The CDR circuit measures the voltage of the input signal and makes a bit-by-bit decision: “1” if the symbol voltage is higher (or equal) than the decision threshold (DT) voltage, and “0” if the voltage is lower than the DT voltage. The decision phase (or timing) is the position within the bit (symbol) time slot at which the voltage measurements take place.
The FEC detection unit 220 detects the ECC, calculates the number of errors (BER), and instructs the control unit 230 to adjust the settings of the CDR to reduce the BER.
Although this technique effectively lowers the BER, it carries a number of drawbacks. For example, it is the characteristic of the receiver which must be adjusted; however, the ECC is generated by the transmitter and the detection unit for interpreting the ECC is typically located outside of the receiver. At best, this may mean that a communications path needs to be provided between the detection unit and the receiver, which increases the response time between interpreting the ECC and applying a change to the receiver characteristic. At worst, the transmitter and detection unit may not be set up to provide FEC at all, and therefore this information is simply not available to the receiver.
Even when such FEC information is available to the receiver, using the FEC error signal can be an inefficient way to adjust receiver settings. For example, if an adjustment is made to a characteristic of the receiver and the BER is observed to decrease, it is possible that the change to the characteristic improved the BER (and hence should be maintained). However, it is also possible that a condition in the transmission line, and not the receiver, changed the BER. Accordingly, the change to the receiver settings may not have affected the BER, or may have actually degraded the signal quality. Additionally, as the BER approaches zero, small changes to the receiver settings may have a disproportionately large effect on the BER, or may have no observable effect at all. In this case, the use of FEC to tune the receiver settings becomes difficult or impossible.
Accordingly, reliance on FEC can be an inefficient and undesirable characteristic of an optical receiver.